Noise in a power supply is rarely a first-order concern for a general-purpose circuit board. For medical instrumentation, that calculus inverts completely. Noise margins that a telecom engineer would shrug at can corrupt an ECG baseline, introduce artifacts into an ultrasound image, or push a photodetector into nonlinearity. The power conversion stage is not a secondary concern, it is frequently the first place a design fails its bio-signal chain.

This article addresses the practical engineering decisions that make a low-noise DC/DC converter suitable, or unsuitable, for clinical-grade instruments.

Why Medical Applications Demand a Different Standard

Medical front-end circuits are almost universally dealing with microvolt- to millivolt-level signals sitting on top of a much larger common-mode voltage. A 12-bit ADC sampling a 5 mV full-scale bio-signal has an LSB size of about 1.2 µV. If the power rail supplying that ADC carries 10 µV of switching ripple at the converter's fundamental frequency, you have already consumed multiple bits of dynamic range before signal processing begins.

The problem compounds with isolation requirements. Patient-connected circuits must meet IEC 60601-1 reinforced isolation thresholds, which typically means a transformer-isolated DC/DC converter is sitting in the power path. Isolated converters introduce a new noise coupling mechanism: capacitive feedthrough across the transformer's interwinding capacitance. A converter with high dV/dt switching transitions pushes common-mode noise across even a well-constructed isolation barrier.

The net result is that noise on a medical power supply is a multi-dimensional problem: there is differential ripple, common-mode conducted noise, and radiated fields that couple directly into analog traces.

Key Specifications to Scrutinize

Output Ripple Voltage

Ripple spec on a data sheet is almost always measured under a specific load and filter condition. Dig into the test circuit. A manufacturer who reports 5 mV ripple on a 5 V output using a 10 µF ceramic capacitor on the output will produce a different result in your design if your output capacitor has a different ESR or if you are operating at light load.

For precision analog applications, target output ripple below 1 mV peak-to-peak under worst-case load. That threshold is achievable with a synchronous topology combined with post-regulation, a low-dropout linear regulator (LDO) after the DC/DC stage trades efficiency for noise performance and is standard practice in high-end signal acquisition hardware.

Switching Frequency and Its Harmonics

The converter's switching frequency and its harmonics will appear in the output spectrum. For ECG applications, the clinical bandwidth runs to 150 Hz; for EEG, to 70 Hz; for EMG, to 10 kHz. A converter running at 100 kHz will not inject fundamental energy into those bands directly, but its harmonics are a different story if the converter's spectral purity is poor or if the layout allows switching noise to couple into analog power planes.

Spread-spectrum frequency modulation, where the switching frequency is dithered over a small range, reduces the peak amplitude of any single tone at the cost of spreading energy across a wider bandwidth. In some medical applications this is beneficial; in others, it complicates EMC pre-compliance testing. Know which mode your application demands before selecting a converter topology.

Isolation Voltage and Test Conditions

IEC 60601-1:2005 (third edition) defines working voltage limits for patient-connected equipment. Converters marketed for medical use should call out whether their isolation rating is an applied test voltage (the 60-second hi-pot value) or a continuous working voltage. These numbers are not the same. A 1000 Vrms hi-pot rating does not imply a 1000 Vrms working voltage, the ratio between test and working voltage is typically 2:1 or higher.

Always verify that the converter manufacturer has conducted or can provide third-party isolation testing data, not just a claims-based datasheet entry.

Topology Choices and Their Noise Implications

Unregulated vs. Regulated Isolated DC/DC

Unregulated isolated DC/DC modules, often driven by a royer oscillator, run at relatively low frequencies (typically 40, 150 kHz) and produce a sinusoidal-like drive waveform. The soft switching reduces high-frequency harmonic content compared to a hard-switched PWM converter. The tradeoff is output impedance: unregulated converters have load-dependent output voltages, making them unsuitable as direct supplies for precision references. They work well as the first isolation stage when followed by a linear post-regulator.

Regulated isolated converters (using PWM control with feedback across the isolation barrier, either via optocoupler or transformer-coupled feedback) offer tighter output regulation but reintroduce the hard-switching noise signature. For medical designs, the preferred architecture is often an unregulated isolator feeding an LDO, with total solution size and cost managed at the system level.

Charge Pump Converters

Charge pumps generate voltage conversion without a magnetic component. This eliminates one significant noise source, the radiating inductor, but introduces charge redistribution transients at the clock frequency. For very low-power analog circuits (sub-1 mA loads), a charge pump can deliver lower integrated noise than an inductor-based solution because the fundamental frequency can be pushed very high (>1 MHz) where filtering is easier.

Layout Practices That Make or Break the Design

A converter's datasheet noise figure is a best-case number measured on a manufacturer's optimized test board. Real designs degrade from that baseline. The most common failure modes:

1. Shared ground planes between the switching converter and the analog signal return. Even a few milliohms of shared impedance injects switching transients into the analog reference.
2. Output filter capacitors placed too far from the load, allowing resonance between trace inductance and the filter capacitance to create a noise peak in the wrong frequency band.
3. Transformer or inductor oriented so the fringe field couples into adjacent analog traces. A 90-degree rotation is often sufficient to reduce pickup by 15, 20 dB.
4. Missing or undersized bypass capacitance on the input side of the converter, allowing input ripple to modulate the converter's switching behavior and create sub-harmonic noise on the output.

Practical Qualification Steps

Before committing a converter to production, run three tests that rarely appear in standard design review checklists:

First, measure output noise with a spectrum analyzer, not an oscilloscope. An oscilloscope's bandwidth-limit filter obscures high-frequency content that still passes through to the load. A spectrum analyzer measurement from DC to 10 MHz gives a full picture of the converter's spectral signature.

Second, test at the extremes of your input voltage range and at 20% and 100% load. Converter noise is often worst at one of these corners, not at the nominal design point.

Third, characterize common-mode noise using a common-mode current probe on the output leads. This quantifies the noise that your isolation barrier is failing to block and is the dominant failure mode in patient-connected circuits.

Closing Thoughts

Selecting a low-noise DC/DC converter for medical instrumentation is an engineering discipline, not a procurement exercise. Ripple voltage, spectral content, isolation quality, and layout sensitivity all interact. An appropriate converter selected carelessly will fail; a conservative one implemented well will meet clinical signal quality requirements with margin. The time spent on power supply characterization is never wasted in this domain.

For further reading, see the companion articles on minimizing EMI in high-performance power conversion systems and high-voltage power supply design for military electronics.

Read application note

Every switching power converter is a conducted and radiated noise source. That is not a flaw in the design, it is an inherent consequence of rapidly switching current through inductive and capacitive elements. The engineering task is not to eliminate switching noise but to contain it below the thresholds set by regulatory standards and, more immediately, below the levels that degrade system performance.

This article covers the mechanisms that generate EMI in switching converters, the filter architectures that suppress it, and the layout and grounding practices that prevent it from defeating your filters in the first place.

The Two EMI Problems Are Not the Same

Conducted EMI travels back through the power input leads and into the source distribution network. Radiated EMI propagates through the air and couples into nearby circuits or antennas. They share root causes but require different mitigation strategies.

For conducted emissions, regulatory limits in CISPR 22 (information technology equipment) and CISPR 11 (industrial equipment) define allowable voltage noise on the AC mains or DC input lines from 150 kHz to 30 MHz. For radiated emissions, limits apply from 30 MHz to 1 GHz or beyond.

Most EMI problems in DC/DC converters show up first in conducted emissions testing, because the switching frequency and its lower harmonics fall squarely in the 150 kHz, 30 MHz conducted band. Fix conducted emissions well and radiated emissions often follow. The reverse is not always true.

Where the Noise Is Generated

Common-Mode vs. Differential-Mode Noise

Differential-mode (DM) noise appears between the positive and negative supply conductors. It is generated by the switching current that flows in the converter's power loop, from input capacitor, through the switch, through the inductor, back to the capacitor. The size of this loop and its associated inductance determine how much voltage is induced across the loop each switching cycle.

Common-mode (CM) noise appears on both supply conductors simultaneously with respect to chassis or earth ground. It is generated primarily by the dV/dt on switching nodes, drain of a MOSFET, or the switching node of an inductor, capacitively coupling through to chassis through stray capacitances in the layout, transformer interwinding capacitance, and heatsink mounting capacitance. CM noise is typically harder to suppress than DM noise because its coupling paths are parasitic and difficult to characterize precisely.

In a practical converter, you always have both. Most EMI filter designs target DM and CM components separately, with X-capacitors handling DM and common-mode chokes with Y-capacitors handling CM.

EMI Filter Architecture

A standard two-stage EMI filter for a DC/DC converter input includes:

1. A common-mode choke (high impedance to CM currents, low impedance to differential supply current)
2. Y-capacitors from each supply rail to chassis (providing a low-impedance return path for CM noise current, keeping it off the supply leads)
3. X-capacitor across the supply rails (reducing DM ripple voltage)
4. A second-stage LC filter if the converter's switching noise is particularly aggressive

The common-mode choke is the most sensitive component to get right. Its CM inductance must be high enough at the converter's fundamental switching frequency to present meaningful impedance, but its leakage inductance (which appears in the differential-mode path) must be controlled. Leakage inductance that is too high creates a series resonance with the X-capacitor that amplifies noise at the resonant frequency.

For a 100 kHz switching converter, a CM choke with 1, 4 mH CM inductance is a typical starting point. Verify that the choke's rated current exceeds your maximum DC operating current with margin, CM chokes saturate, and a saturated choke provides no EMI attenuation at all.

Layout: Where Most Filters Fail in Practice

An EMI filter that looks correct on a schematic can be completely ineffective on a PCB. The three most common failure modes are:

Input and Output Traces Routed in Parallel

If the converter's noisy output traces run parallel to and close to the filtered input traces, the noise couples directly from output to input by mutual inductance and capacitance, bypassing the filter entirely. The fix is physical separation and orthogonal routing where separation is not possible.

Y-Capacitors Without a Low-Impedance Chassis Connection

Y-capacitors divert CM noise current to chassis ground. If the chassis connection is made through a long PCB trace or a wire with significant inductance, the connection is not low-impedance at the frequencies of interest. At 10 MHz, 10 nH of inductance has an impedance of 628 mΩ, which is significant compared to a 1 nF Y-capacitor's impedance of 16 Ω at the same frequency. Use multiple vias in parallel to chassis ground connections and keep traces short.

Filter Components Placed After Rather Than Before the Noisy Switching Node

Power converter PCB layout must treat the EMI filter as a clean-to-noisy boundary. Components on the input side of the filter should be physically separated from the switching components of the converter. Any trace that connects across this boundary without going through the filter defeats its purpose. This means the input filter capacitor must be located close to the input pins of the converter module, not close to the external filter.

Spread-Spectrum Clocking: Benefits and Limitations

Spread-spectrum frequency modulation (SSFM) reduces the peak spectral amplitude of switching harmonics by distributing energy across a band rather than concentrating it at a single frequency. A converter that produces a 60 dBµV peak at 100 kHz with a fixed clock might produce a 50 dBµV peak spread across 90, 110 kHz with SSFM, a 10 dB reduction in peak amplitude.

SSFM is genuinely useful for approaching conducted emissions limits when the fundamental or a lower harmonic is failing by a small margin. It is not a substitute for good filtering. If you are 20 dB over the limit, SSFM will not save you. It also complicates synchronization with other clocks in the system, and some applications (medical, precision timing) cannot tolerate a varying switching frequency.

Ferrite Beads as a Supplemental Tool

Ferrite beads are resistive at their target frequency range, not reactive. A bead specified at 600 Ω at 100 MHz provides that impedance through loss, not inductance. This makes them effective for high-frequency conducted noise (above approximately 10 MHz) where they absorb energy rather than reflecting it back to the source.

Do not use a ferrite bead as a primary EMI filter element at the converter's switching frequency. At 100, 500 kHz, most ferrite beads are still predominantly inductive and provide little attenuation. Their value is in suppressing high-order harmonics and fast-edge transients, not the fundamental.

Measurement Before Commitment

Pre-compliance testing with a LISN (line impedance stabilization network) and a spectrum analyzer is strongly recommended before final filter design commitment. Real EMI measurement requires a defined source impedance for the input side, which the LISN provides. Without it, your filter response is dependent on the source impedance of your bench supply, which is not the same as the impedance seen in end use.

A two-day pre-compliance test session before final PCB spin is far cheaper than a failed full compliance test and a board respin.

Closing Thoughts

EMI management in power conversion is a systems discipline. The filter, the layout, the converter topology, the chassis connection, and the grounding architecture all interact. Treating any one of them in isolation produces a design that is full of surprises on the test bench. Get the full picture first.

For related topics, see the articles on low-noise DC/DC converters in precision medical instrumentation and high-voltage power supply design for military electronics.

Read application note

Designing power supplies for military applications is not the same discipline as consumer or industrial power supply engineering with a few extra checkboxes added. The environmental envelope, the reliability requirements, the documentation burden, and the electrical specifications all impose constraints that fundamentally shape design choices from the beginning. High-voltage supplies in particular combine the stress management challenges of any HV design with the operational demands of MIL-SPEC environments.

This article addresses the specific engineering considerations that separate a successful military high-voltage power supply from one that fails acceptance testing or field deployment.

Defining "High Voltage" in Military Context

In military electronics, high-voltage power supplies typically refer to outputs above 500 V DC, though many defense applications need supplies in the 1 kV to 30 kV range. Applications include:

• Traveling wave tube amplifiers (TWTAs) in radar and electronic warfare systems requiring 3, 8 kV collector voltages
• Geiger-Müller and scintillation detector bias in nuclear detection systems (typically 500, 2000 V)
• Image intensifier tube supplies in night vision equipment (typically 600, 1200 V)
• Laser driver supplies for rangefinders and designators
• Klystron and magnetron supplies in high-power radar transmitters

Each application has distinct load characteristics, TWTAs are current-limited, detector bias supplies must be noise-minimal, laser drivers are pulsed, and that directly affects topology selection.

Environmental Standards and What They Actually Require

MIL-STD-461 governs electromagnetic emissions and susceptibility for military equipment. For power supplies, the critical requirements are CE101 (conducted emissions on power leads, 30 Hz to 10 kHz), CE102 (conducted emissions on power leads, 10 kHz to 10 MHz), and CS101 (conducted susceptibility, the supply must continue operating correctly in the presence of injected low-frequency noise on its input).

CS101 testing injects a swept sinusoidal signal at up to 30 V amplitude across the input leads. A supply with a poorly designed input filter may oscillate or shut down when this signal interacts with the filter's resonance. The fix is damping the filter's Q, either by adding series resistance (inefficient) or by designing the filter with sufficient intrinsic damping. This is not obvious from a schematic review alone; it requires small-signal stability analysis of the input filter and converter together.

MIL-STD-810 covers environmental factors: temperature, altitude, humidity, vibration, and shock. For a high-voltage supply, the most critical environmental parameter is often altitude. Reduced air pressure reduces the dielectric strength of air gaps and the effectiveness of convective cooling simultaneously. At 70,000 feet (roughly 21 km), atmospheric pressure is approximately 6% of sea level. Creepage and clearance distances that are adequate at sea level are completely inadequate at altitude. Designs for airborne applications must either be fully potted (hermetically sealing out air) or engineered to the reduced-pressure dielectric requirements explicitly.

MIL-PRF-28748 and related performance specifications define qualification testing procedures for military power supplies including output regulation, transient response, and insulation resistance.

Topology Considerations for High-Voltage Outputs

Flyback Converter

For outputs up to approximately 3 kV at lower power levels (sub-50 W), the flyback topology is common because it requires only a single active switch and the transformer provides inherent isolation. High turns-ratio transformers for HV flybacks must be carefully wound to control leakage inductance, which creates voltage spikes on the primary switch at turn-off. In HV designs, an uncontrolled spike can exceed the MOSFET's breakdown voltage, standard RC or TVS clamp networks must be designed precisely.

Resonant Converters

For higher power HV applications, LLC resonant converters offer soft-switching advantages. Zero-voltage switching (ZVS) on the primary and zero-current switching (ZCS) on the secondary reduce switching losses and, importantly, reduce the dV/dt of switching transitions. Lower dV/dt means reduced EMI and reduced stress on transformer insulation. LLC converters require more complex control and have limited range of output voltage regulation, often necessitating a second stage for output voltage adjustment.

Voltage Multiplier (Cockcroft-Walton) Ladders

For very high voltages (10 kV and above) at modest current, Cockcroft-Walton multiplier ladders driven by a resonant converter are practical. The multiplier is passive (diodes and capacitors only), which simplifies insulation management, each stage only supports a fraction of the total output voltage. The penalties are energy storage capacity (poor transient response) and the requirement for tight regulation of the driving AC source.

Insulation and Creepage in Practice

The IPC-2221 standard and IEC 60664-1 provide creepage and clearance tables, but for military HV designs the designer should use them as starting points, not as gospel. Both standards assume defined pollution degrees and overvoltage categories. Military designs operating in unpredictable environments should assume worst-case contamination (Pollution Degree 3 in IEC 60664 terms) and design creepage distances accordingly.

For a 3 kV DC output operating in a Pollution Degree 3 environment, IEC 60664-1 requires approximately 8 mm creepage distance for basic insulation. For reinforced insulation (the category required when HV could contact accessible conductive parts), that doubles to 16 mm. On a compact PCB, achieving 16 mm of creepage between a 3 kV trace and any other conductor requires either aggressive trace routing or a physical barrier such as a slot cut through the board material.

Potting compounds solve many HV insulation problems by filling air gaps. Epoxy, polyurethane, and silicone potting each have different thermal conductivity, flexibility, and outgassing characteristics. Silicone potting is preferred in applications with wide temperature cycling because its flexibility prevents it from cracking away from potted components and creating voids, voids under HV stress are a failure mode called partial discharge that degrades insulation over time without immediate visible failure.

Output Regulation and Noise for Detector Applications

High-voltage bias supplies for particle detectors and photomultiplier tubes impose exceptionally tight regulation and noise requirements. The detector's energy resolution is directly proportional to the stability of its bias voltage. A 0.01% change in PMT bias can shift the gain by 0.5, 1%, corrupting energy calibration in nuclear detection equipment.

For these applications, the output ripple must typically be below 10 mV peak-to-peak on a 1000 V supply (10 ppm). Achieving this requires:

1. A precision high-voltage reference divider in the feedback loop, using high-stability resistors (25 ppm/°C or better) with matched temperature coefficients
2. An output filter capacitor selected for stability over temperature and voltage, not all HV capacitors maintain capacitance well at their rated voltage
3. Post-regulation using a high-voltage series-pass element (depletion-mode JFET or high-voltage BJT) to suppress residual switching ripple

Reliability and Derating

Military supplies are expected to operate reliably over 10,000+ hour lifetimes in harsh environments. Component derating is mandatory: capacitors operated at 50% of rated voltage, MOSFETs at 70% of VDSS, resistors at 50% of rated power. For HV circuits, capacitor derating is especially important because electrolytic and ceramic capacitors both suffer accelerated aging at high electric field stress.

MIL-HDBK-217 provides failure rate prediction models for electronic components. Though the model is dated and controversial among reliability engineers, many defense procurement programs still require 217-based MTBF calculations, and a design that derates components correctly will always produce better MTBF predictions, as well as better actual reliability.

Closing Thoughts

Military high-voltage power supply design rewards engineers who engage with the standards rather than treating them as compliance hurdles. The requirements for EMC, environmental survivability, insulation coordination, and reliability derating are not arbitrary, they reflect hard lessons from field failures in demanding environments. Starting with those requirements and working backward into the design is more efficient than designing conventionally and attempting to retrofit compliance.

For complementary topics, see the articles on low-noise DC/DC converters in precision medical instrumentation and minimizing EMI in high-performance power conversion systems.

Read application note